FIG. 1a shows a power supply system for a typical battery-powered peripheral device.
The system comprises connections 11 and 13 to external power supplies, as well as to a re-chargeable battery 10. The external supplies are typically regulated (12) and/or switched to a common external supply node (Vsup) which supplies charging current to the battery through a charging circuit or controller 14. The battery 10 is coupled to an internal power bus which supplies one or more load regulators 15 which provide regulated voltage outputs to one or more parts of the peripheral device. For example one load regulator 15a may supply the disk drive of an MP3 player, whilst another regulator 15b supplies its signal processing and amplifier circuitry.
The battery 10 can be recharged from the power wires 11 in a bus cable, such as a USB or IEEE 1394 connection. Supply current taken from the bus power wires 11 will generally first pass through a supply bus regulation block 12. In the case of USB, this is required to guarantee that the current taken from the bus be limited to 100 mA or 500 mA. For the case of 1394 the supply regulator 12 is required to attenuate a possible 48V to the 5V or so maximum that the power supply system circuitry can tolerate. Techniques for such regulators, generally involving sensing the output voltage and current and feeding these signals into one or more feedback loops are well known to those versed in the art.
In an alternate mode of operation, some or all of the required supply current can be sourced externally (13) via a transformer (not shown) attached to the mains, or perhaps from a 12V nominal source from a car battery. This supply voltage is again generally pre-regulated to 5V or so, for example by a linear or switching regulator (not shown). There may also be a means for selection between the bus (11) and the external (13) supplies, for example either diodes in series with one or more of these supplies or more intelligent control with comparators and controlled switches. For simplicity these are not shown in FIG. 1. These non-battery supplies (11, 13) are coupled to a common node or voltage rail Vsup, which is then supplied by one of these external supply sources (11 or 13).
Current supplied to the battery 10 must be regulated in current to limit the current when charging, and in voltage to prevent over-charging of the battery. This function is achieved using a charging control block or circuit 14. For example a Li-ion battery will typically be charged at constant current (typically 0.5 to 1.0 C amperes, where C is the battery capacity in ampere-hours, say 800 mAh) until its terminal voltage reaches 4.2V, then it will be charged at constant voltage of 4.2V until the current taken drops to near zero. Techniques for such charger regulators 14, generally involving sensing the output voltage and current and feeding these signals into one or more feedback loops, are well known to those versed in the art.
Depending on the state of charge of the battery 10, its output voltage Vbat can vary from say 4.2V when fully charged to as low as about 2.7V before the battery becomes so discharged as to cause irreversible degradation in its capacity. As high-current loads (e.g. motors) are switched on and off, this battery terminal voltage may also vary due to the output impedance of the battery (say 100 mΩ). Some circuitry may be able to accept this unregulated voltage direct from the battery. This may be attractive in terms of system cost. However most circuitry will require a cleaner, better regulated, supply, perhaps regulated at 1.8V or 3.3V for logic circuitry, and higher voltages for other applications, such as 7.2V for driving banks of white LEDs for example. There will thus generally be one or more voltage regulators 15a and 15b driven from the battery line Vbat.
Depending on the input and output voltage levels, the efficiency required, and the cleanliness required of the supply, these voltage regulators 15 may be capacitive charge pumps, or inductive buck or boost switching regulators, or linear regulators. These are shown in simplified from in FIG. 1a, and in more detail in FIG. 2. FIG. 2a shows a low-drop-out linear regulator; FIG. 2b shows a non-low-dropout linear regulator; FIG. 2c shows a buck switching regulator; FIG. 2d shows a buck-boost switching regulator; FIG. 2e shows a boost switching regulator, FIG. 2f shows a non-inverting boosting capacitor charge pump regulator, and FIG. 2g shows an inverting capacitor charge pump regulator. Many other well-known variants of regulator exist, including those where diodes are replaced by appropriately switched pass transistors.
Except for the simple boost regulator (FIG. 2e), all these circuits contain a switch-type input pass device Mp connected directly to the input supply, and to a regulator internal or output node Vx. The charge pump of FIG. 2f includes two such devices. The boost regulator of FIG. 2e includes an input inductor rather than a switch-type device and which is connected to a similar internal node Vx.
However boost regulators will typically incorporate a current sense resistor or MOS mirror arrangement to sense the input current to give better loop stability. A MOS mirror arrangement coupled to the input of a boost regulator is shown in FIG. 2h. This includes a pass device Mp inserted in series with the input supply Vdd, and a smaller MOS mirror device Mps connected in parallel, with common source and gate connections. This generates a scaled replica of the current through Mp, which can help stabilise the PWM control loop. The current monitoring function can also be useful in implementing a current limit function to protect the circuit. The pass device Mp can also be used to isolate the battery from the output to prevent leakage when the switcher is off; for example from a battery to an output load.
Note that the various pass devices are shown as MOSFETs but may be any suitable device including NMOS, PMOS, diodes, or bipolar transistors or even relays where suitable.
Generally, if the alternate supply (13) is available, it will be used in preference to the bus supply (11) or the battery (10). If no alternate supply is available, the bus supply will be used if possible. Only if neither the bus supply nor the alternate supply is available will the battery supply (10) be used. This operation can be realised for example by sensing the voltage on the various supplies and controlling various switches depending on which of these supplies exceed respective thresholds. Such control techniques are well known to those skilled in the art.
An example of a similar type of power supply is disclosed in Maxim Integrated Products' data sheet reference MAX1874. As shown in FIG. 1b, this merges transistors or pass devices Ma and Mb and their controls 12 and 14 from FIG. 1a, and couples the alternate supply to the battery via a parallel transistor Mb2 and control 14′. This chip does not include the downstream regulators, but they would typically be connected to the battery as shown, to allow the system to function powered from the battery in the absence of the supplies.
One problem with this type of scheme relates to the time which the system takes to become active when powered from the bus (11) or alternate (13) supply with a discharged battery 10. The load regulators 15 will have a minimum input voltage, perhaps 3.2V, (or maybe as high as 3.6V for a 3.3V linear low-dropout regulator—FIG. 2a), whereas the battery may initially be discharged below this voltage. Thus the system supplied by the load regulators 15 will not work properly until the battery 10 is charged up. Where the battery is heavily discharged, this might take several minutes or longer. If the battery is deeply discharged, below say 2.5V, the battery charging current is in fact typically reduced by a factor of ten in order to minimize battery capacity degradation effects and also as a safety mechanism due to the fact that in the absence of adequate power there may be no software control of the system. In this situation the wake-up time will clearly be even longer. This behaviour is undesirable to consumers who now desire “instant-on” behaviour.
A further problem is that the charger current control 14 or 14′ limits the current to the node Vbat, to avoid too rapid charging of the battery 10. However it cannot differentiate between current taken by the battery 10 and that taken by the other loads, for example the regulators 15. Thus if the battery charging current is limited to 100 mA, then the total taken by the loads is also limited to 100 mA. Thus if they take 99 mA, only 1 mA is available to charge the battery, further increasing the time required for the system to operate properly. Even if the error is less gross than this, and say there is only a 25% reduction in charger current actually reaching the battery, this may well confuse the analog or digital control of the battery charging process, affecting the effective Icharge-Vbat trajectory, and causing a charging time that is still sub-optimum, even allowing for the 25% reduction in battery charging current.
In this kind of scheme the system current is also limited to the maximum current allowed by the charger 14 or 14′, which means that whenever the overall system current (including regulator input current) requires a higher current than allowed by charger control, this current would be drained out of the battery 10. This is not just extending charging time it is also decreasing the battery life time.
The circuit could be improved by sensing current flowing only into the battery 10, while controlling all current into Vbat, but this still does not guarantee adequate current into the other loads 15, as the splitting of current will be defined by the respective V-I characteristics of all circuits connected to the Vbat node, including the battery, so a discharged battery would tend to steal current away from loads expecting a higher voltage. This means that this current would be taken by the battery 10 as a priority, rather than by the loads 15 as a priority
FIG. 3 illustrates one solution to this problem of the “instant-on” requirement. The load regulators 15 are now supplied from the bus supply and external supply common node Vsup, rather than directly from the battery 10. As the battery node Vbat can be isolated (switch Mc and charging control 14) from this common supply node Vsup, the system can wake up as soon as power is applied either from the bus (11) or the alternate (13) supply. Only when neither the alternate supply (13) nor the bus (11) can supply current, an additional battery switch Mc is turned on, and the load regulators 15 are then supplied from the battery 10.
Examples of similar arrangements are disclosed in Linear Technology Corporation's data sheet references LTC3455 and LTC4055.
In these cases the battery charger supplies only the battery, so the charging current can be accurately monitored to allow intelligent control of the charging current-voltage trajectory.
Also when driving the system from the bus or alternate supply, this arrangement avoids the power losses associated with passing current through the charger regulator prior to being input to following switching regulators. Efficiency per se may not be a major concern when driving from non-battery supplies, but reducing power dissipated may allow less heat-sinking and hence lower system cost.
The main problem with this solution is the extra voltage drop between Vbat and Vsup when the load regulators 15 are driven from the battery 10 compared to the system of FIG. 1. The battery voltage is at best 4.2V, and should work down to as low as possible to extend operating time between battery recharging (albeit avoiding deep discharge, below about 2.6V). The load regulators 15 require a minimum input voltage (regulated output voltage plus dropout voltage) in order to maintain regulation of their output voltage, so the regulators will continue to function correctly until the battery discharges to this minimum input voltage. However the voltage drop across this additional switch device (Mc) effectively increases the minimum voltage required from the battery, and hence reduces the time the battery can provide this. The voltage drops across switch devices (Ma, Mb, Mc) increases with their on-resistance.
Given the technologies available today, these switch devices will generally be implemented using MOS switches, rather than bipolar transistors or relays. Lower on-resistance discrete MOS switches are more expensive as they require larger silicon area or more complex and specialized wafer processing. More particularly, for systems where most of the circuitry of FIG. 1 or 3 is implemented on a single chip, the total area required for these switches has not only an impact on chip area and hence cost, but also may require so much area that the silicon die may not fit in the desired plastic package. This is especially critical for portable equipment such as MP3 players or mobile phones, where the size of the whole system is an important specification and requires the smallest possible package size.
To allow Vbat to reach 4.2V when fed from a 4.5V bus supply (11), Ma and Mb might be sized to drop 150 mV each at peak battery charging current. But the sizes of input transistors Mp in the switching or linear load regulators 15 will define a minimum input voltage to keep their respective outputs in regulation. So either a substantial reduction in operating battery life has to be tolerated, or the input switches Mp of the load regulators 15 have to be greatly enlarged and possibly even extra bond wires and package terminals added as the parasitic resistances involved in tracking the current from chip to the outside world are significant. For example if a minimum battery voltage of 3.6V has to supply a 3.3V output linear dropout regulator 15b, then battery switch Mc and the regulator's pass device Mp have to be designed for a 150 mV drop-out voltage each at peak load current.
There is also a possible issue of problems arising from modulation of the voltage on Vsup caused by load variations on the load regulators 15. As downstream peripherals are plugged in, or as a disc drive internal to the battery-powered peripheral starts up, there can be a rapid surge in supply demanded from one regulator 15a. This will appear as a current step on Vsup, giving a voltage step across the on-resistance of Mc, and this may be enough to transiently reduce Vsup below the minimum input voltage for another regulator 15b on Vsup, or at best give a transient on this regulator output due to its finite line regulation. Even when the line regulation is good at d.c., it falls off with frequency, so voltage steps on Vsup may still give transients on the regulator outputs.
If Mc is controlled in a local regulation loop, rather than just being turned on, this may reduce transients, but this loop will again have finite gain and bandwidth, so there will still be transients at some level. This would also increase the complexity and hence cost of the circuit arrangement. Also if Mc is regulated for example to deliver a Vsup at a fixed voltage difference below Vbat, this voltage difference will then have to be set to a worst-case voltage drop, which will make battery voltage headroom under non-maximum load conditions even worse.